N/A
The fields of power electronics, and power supplies in general, are concerned with the processing of electrical power using electronic devices. One class of power supply that is commonly used to provide power for electronic devices such as personal computers, laptop computers, personal communication devices, and personal digital assistants is referred to as a DC/DC switching converter power supply. In general, a DC/DC switching converter power supply contains a raw power input port that is typically coupled to a DC power source such as a battery and controller. The raw DC input power is processed according to one or more control signals provided by the controller and yields a conditioned output power signal. In particular, a DC/DC converter converts a DC input voltage to a conditioned DC output voltage that may have a larger or smaller voltage magnitude. One type of DC/DC converter is a forward converter illustrated in FIG. 1. Typically, a DC/DC forward converter is used to provide a DC output voltage that has an output magnitude less than the input magnitude. In particular, as depicted in FIG. 1, a typical prior art forward converter 100 includes a raw DC voltage source 107 input between terminals 101 and 105 and coupled to the primary winding 103 of a power transformer 102. A transformer reset circuit 104 is provided to demagnetize the power transformer during periods when no current is present in the primary coil. A first switch 106, which is typically a power switching MOS transistor, is coupled between the primary winding of the power transformer 102 and the reference terminal of the voltage input 105. The secondary coil 109 of the power transformer 102 is coupled to switching diodes 110 and 112, inductor 114, and output capacitor 116, wherein zn output voltage is developed between terminals 117 and 119. If the first electronic switching module is not a power MOSFET switch a protection diode 108 may be placed across the first electronic switching module to provide a discharge path for the inductance in series around the switching module 106. A power MOSFET does not need a protection diode due to a parasitic diode that is inherently created due to the semiconductor structure of the MOSFET.
When an input pulse is applied to the primary winding 103 of the transformer 102 a voltage is induced in the secondary winding 109 of the power transformer 102, the polarity of which is indicated by the respective dots shown on the windings in FIG. 1. Accordingly, during a positive going pulse, switching diode 110 turns on and a circuit is formed that includes the secondary winding 109 of transformer 102, inductor 114, capacitor 116 and switching diode 110. During the positive going pulse, inductor current IL 120 flows into the inductor from the secondary winding. The inductor current 120 is equal to the integral of the voltage applied to the inductor divided by the inductance thereof. Accordingly, for a square wave pulse having a constant amplitude, the inductor current 120 will begin to increase in a substantially ramp like manner.
Similarly, when the first switching module 106 turns off the input voltage pulse, switching diode 110 turns off and the inductor current 120 begins to decrease as a linear function. Switching diode 112 will turn on when the voltage at node 121 has fallen below the threshold voltage of diode 112. When conducting, switching diode 112 turns on to complete the circuit that includes switching diode 112, inductor 114 and output capacitor 116.
The output voltage provided between output terminals 117 and 119 is a function of the amplitude of the voltage input pulses provided by the input DC voltage source 107, the turns ratio of power transformer 102, the switching frequency of the first switching module 106, and the duty cycle of the input pulses. For the forward converter illustrated in FIG. 1, the output voltage is less than or equal to the voltage across the secondary winding of the power transformer.
The switching diodes 112 and 110 each have a small non-zero resistance when forward biased, i.e., when the diodes are turned on and are conducting. When any current is flowing through the respective diode, the non-zero resistance results in a voltage drop being generated across the switching diode, resulting in a diode conduction loss equal to V*I, where V is the voltage drop and I is the current flowing through the diode. For a typical switching Schottky diode, this diode voltage drop may be as high as 0.4 volts. Because the switching diodes 110 and 112 are in each of the two circuit paths, the output voltage, which is less than the input voltage to begin with, is further reduced by the diode voltage drop. In some DC/DC converters, the diode conduction loss can contribute significantly to the overall power loss. In low output voltage applications the diode conduction loss can be particularly serious. Thus, as the supply voltages for next generation electronic equipment become lower, the forward conduction loss of the switching diodes becomes increasingly significant.
Many components used in current electronic products require 3.3 volts, and in some cases, most notably microprocessors, the voltage requirements have dropped below 2 volts. As this trend of lower supply voltages continues into the future, many electronic devices will be designed to operate at 1 volt or less. As an example of the problems associated with Schottky switching diodes in DC/DC power supplies, a power supply having an output voltage of 5 volts will have approximately 92% to 93% efficiency. However, as the output voltage drops the efficiency of the diode rectifiers drops as well. At 3.3 volts for example, the efficiency of the diode rectifiers is approximately 88%, at 2 volts the efficiency is approximately 83%, and at 1 volt the efficiency is less than 75%.
To ameliorate this condition, the switching diodes 110 and 112 used in FIG. 1 are often replaced with other electronic switching modules that may include single or multiple MOSFETs, bi-polar transistors, or other semiconductor switches such as thyristors or SCR""s. Typically, these electronic switches are referred to as synchronous rectifiers since they are switched on and off synchronously with the switching cycles of the first switching mode to rectify the pulsed DC voltages induced in the secondary coil 109. Typically, synchronous rectifiers are large channel area power MOSFET switches that are able to clamp the various switching nodes to 0.1 volt or less thus reducing the forward conduction loss by a factor of 4 or more when compared to Schottky switching diodes. Synchronous rectifiers are typically driven using one of two methods. In the first method, a control circuit is used to drive the synchronous rectifiers. In this case, the trade off in using a switching diode or a MOSFET rectifier is whether the power needed to drive the MOSFET gate cancels the efficiency gained from a reduced forward voltage drop.
FIG. 2 depicts the second method for driving a pair of synchronous rectifiers in which the synchronous rectifiers are self-driven. A self-driven system does not suffer from the energy losses described above since the energy necessary to drive the gates of the two synchronous MOSFET rectifiers is returned to the inductor or transformer. In particular, a forward converter 200 uses MOSFET switches 210 and 214 to provide the necessary current paths to rectify the incoming power pulses. The efficiency gain of the synchronous rectifiers depends on the load current, the input battery voltage, the desired output voltage, the switching frequency of the first switching module, and the characteristics of the MOSFET switches. Typically in a DC/DC converter, a lower output voltage and higher load current will militate toward the use of synchronous rectification.
Synchronous rectification using MOSFET devices, however, is not without problems. In particular, MOSFET synchronous rectifiers have two disadvantages that decrease the overall efficiency of the power supply. The first problem associated with the MOSFET synchronous rectifiers is known as the reverse recovery condition. The construction of a MOSFET transistor results in a parasitic PN junction between the source-channel-drain regions and the body of the MOSFET. This parasitic PN junction forms a body diode that conducts current between the MOSFET structures and the body during the reverse recovery period when the voltage on the drain has been reduced and the parasitic diode is forward biased. This body diode conduction can result in significant power loss as described above with respect to the synchronous rectifiers used in the forward converter depicted in FIG. 2. Reverse recovery of a MOSFET switch occurs due to the stored junction charge of the MOSFET body diode caused by the current flowing therethrough. Because this junction charge cannot be removed instantaneously, the anode to cathode voltage will remain constant as the MOSFET body diode is switched from forward to reverse bias. At the time the switch occurs, the current through the junction reverses direction and stays at a constant level for a period commonly referred to as the storage time, Dx. Physically, the storage time is the time it takes the electrons to move from the P-material back to the N-material and for the holes to move from the N-material to the P-material is determined by the geometry of the junction. During this period the MOSFET is essentially a short circuit. After the storage time has elapsed, the body diode will turn off and the current will then decrease to the reverse leakage current value of the MOSFET. The time for the current to decrease to the reverse leakage current is commonly referred to as the transition time and the sum of the storage time and the transition time is the reverse recovery time. Physically, the transition time is the time required for the electrons to recombine at the anode, and the holes to recombine at the cathode until there are no more of the original stored carriers left. The transition time of the junction is a function of both the geometry and the doping levels of the junction.
The forward converter 200 depicted in FIG. 2 utilizes a self-drive method for the synchronous rectifiers, in which the gates are coupled to one terminal or the other of the secondary coil 205 of the transformer 202. Because of the inductors and capacitors in the power supply, the voltages and currents do not change instantaneously. As a result of the inherent dynamics of the system, both MOSFET switches 210 and 214 can be conducting simultaneously, wherein one MOSFET switch is turned on and conducting, and the other MOSFET switch is in the reverse recovery period and the body diode and channel are both conducting current.
Another issue with a self-driven synchronous system is the variation in the channel resistance of the MOSFET due to voltage variations in the VGS voltage. The varying channel resistance results in a changing loss and a variation in the output current and voltage.
Therefore, it would be advantageous to provide a control system for a DC/DC converter that minimizes the reverse recovery and parasitic body diode of the synchronous rectifiers used therein.
A control system for controlling a switched mode power supply including first and second synchronous rectifiers is disclosed. The control system provides control signals to each of the synchronous rectifiers such that the body diode conduction of the synchronous rectifier being switched is minimized. The control system achieves this by shifting in time the portion of the control signal turning on the first synchronous rectifier, i.e., the rising edge of the first control pulse, such that the body diode is not forward biased, and hence does not conduct. Similarly, the control system provides the control signals necessary to turn-off the first synchronous rectifier by shifting in time the portion of the control signal turning off the first synchronous rectifier, i.e., the trailing edge of the first control pulse, such that the forward biasing of the body diode is minimized. The control system controls the turn-on of the second synchronous rectifier by detecting the conduction of the body diode of the second synchronous rectifier and adjusting the turn-on signal to minimize or eliminate this conduction period. The control system controls the turn-off of the second synchronous rectifier in a similar manner to the first synchronous rectifier. In particular, the control system shifts in time the turn-off signal to the second synchronous rectifier to coincide with the turning-on of the first synchronous rectifier. This avoids allowing the body diode of the second synchronous rectifier to conduct simultaneously with the conduction of the first synchronous rectifier.
In one embodiment the first and second synchronous rectifiers each have first, second, and control electrodes, and the switched mode power supply includes a pulse width modulator (PWM) providing a plurality of PWM signal pulses each PWM signal pulse corresponding to a switching cycle. The control system includes a first control module that is configured and arranged to provide a first control signal having a turn-on portion and a turn-off portion to the control electrode of the first synchronous rectifier. The first control module receives three input signals: the first input is a first measurement signal indicative of the voltage magnitude between the first and third terminals of the first synchronous rectifier. The second input is a second measurement signal indicative of the magnitude of the first control signal, and the third input is the PWM signal pulse corresponding to the current switching cycle. The first control module is configured and arranged to predict the optimal turn-on time of the first rectifier as a function of the first and second measurement signals of the previous switching cycle, and the pulse signal of the current switching cycle. The first control module is configured and arranged to provide the turn-on portion of the first control signal to the control terminal of the first synchronous rectifier. Similarly, the first control module is further operative to predict the optimal turn-off time for the first synchronous rectifier as a function of the previous switching cycle, and the pulse signal of the current switching cycle. The first control module is configured and arranged to provide the turn-off portion of the first control signal to the control terminal of the first synchronous rectifier. In this way the conduction of the parasitic body diode of the second synchronous rectifier is substantially minimized.
The controller includes a second control module to provide a second control signal to the control electrode of the second synchronous rectifier. The second control module receives a third measurement signal indicative of the voltage between the first and third terminals of the first synchronous rectifier, a pulse signal indicative of the PWM signal corresponding to the current switching cycle, and the first measurement signal. The first control module is configured and arranged to predict the optimal turn-off time for the second synchronous rectifier as a function of the first, second and third measurement signals of the previous switching cycle and the pulse signal of the current switching cycle. The second control module is operative to provide the turn-off portion of the second control signal to the control terminal of the second synchronous rectifier. The second control module is further configured and arranged to predict the optimal turn-on time for the second synchronous detector by detecting when the second measurement signal has first increased above a first threshold level and has subsequently decreased beneath a second threshold level. The second control module is operative to provide the turn-on portion of the second control signal to the control terminal of the second synchronous rectifier, minimizing the conduction of the parasitic body diode of the second synchronous rectifier.
Other forms, features, and aspects of the above-described methods and system are described in the detailed description that follows.